Ejemplo del 741

Ejemplo del 741
Diagrama electrónico del operacional 741.
En el diagrama se destaca en azul el amplificador diferencial. Éste es el responsable de
que las corrientes de entrada no sean cero, pero si muy bajas respecto a las de los
colectores (Nótese como a pesar de aproximar las corrientes de entrada a 0, si éstas
realmente fueran 0 el circuito no funcionaría). La impedancia de entrada es de unos
Though designs vary between products and manufacturers, all op-amps have basically
the same internal structure, which consists of three stages:
1. Differential amplifier
o Input stage — provides low noise amplification, high input impedance,
usually a differential output
2. Voltage amplifier
o Provides high voltage gain, a single-pole frequency roll-off, usually
single-ended output
3. Output amplifier
o Output stage — provides high current driving capability, low output
impedance, current limiting and short circuit protection circuitry
Las etapas en rojo son espejos de corriente. El superior de la izquierda sirve para poder
soportar grandes tensiones en modo común en la entrada. El superior de la derecha
proporciona una corriente a la circuitería de salida para mantener la tensión. El inferior
tiene una baja corriente de colector debido a las resistencias de 5kΩ. Se usa como
conexión de gran impedancia a la alimentación negativa para poder tener una tensión de
referencia sin que haya efecto de carga en el circuito de entrada.
Los pines llamados Offset null son usados para eliminar las tensiones de offset que
pueda haber en el circuito.
La etapa de ganancia en tensión es NPN.
La sección verde es un desplazador de tensión. Esto proporciona una caída de tensión
constante sin importar la alimentación. En el ejemplo 1V. Esto sirve para prevenir la
El condensador se usa como parte de un filtro paso bajo para reducir la frecuencia y
prevenir que el A.O oscile.
La salida en celeste es un amplificador PNP seguidor con emisor push-pull. El rango de
la tensión de salida es de un voltio menos a la alimentación, la tensión colector-emisor
de los transistores de salida nunca puede ser totalmente cero. Las resistencias de salida
hacen que la corriente de salida esté limitada a unos 25mA. La resistencia de salida no
es cero, pero con realimentación negativa se aproxima.
Current mirrors
The sections outlined in red are current mirrors. The primary current, from which other
standing (bias) currents are generated, is determined by the chip's power supply and the
39 kΩ resistor acting (with the two transistor diode junctions) as a current source. The
current generated is approximately (VS+ − VS− − 2Vbe)/39 kΩ. The input stage DC
conditions are controlled by the two current mirrors on the left. The current mirror
formed by Q8/Q9 allows for large common mode voltages on the inputs without
exceeding the active range of any transistor in the circuit. The current mirror Q10/Q11
is used, indirectly, to set the input stage current. This current is set by the 5 kΩ resistor.
The input stage bias control acts in the following manner. The outputs of current
mirrors, Q8/Q9 and Q10/Q11 together form a high impedance current differencing
circuit. If the input stage current tends to deviate (as detected by Q8) from that set by
Q10, this is mirrored in Q9 and any change in this current is corrected by altering the
voltage at the bases of Q3 and Q4. Thus the input stage DC conditions are stabilised by
a high gain negative feedback system.
The top-right current mirror Q12/Q13 provides a constant current load for the class A
gain stage, via the collector of Q13, that is largely independent of the output voltage.
Differential input stage
The blue outlined section is a differential amplifier. Q1 and Q2 are input emitter
followers and together with the common base pair Q3 and Q4 form the differential input
stage. In addition, Q3 and Q4 also act as level shifters and provide voltage gain to drive
the class A amplifier. They also help to increase the reverse Vbe rating on the input
The differential amplifier formed by Q1 - Q4 drives a current mirror active load formed
by transistors Q5 - Q7. Q7 increases the accuracy of the current mirror by decreasing
the amount of signal current required from Q3 to drive the bases of Q5 and Q6. This
current mirror provides differential to single ended conversion as follows:
The signal current of Q3 is the input to the current mirror while the output of the mirror
(the collector of Q6) is connected to the collector of Q4. Here, the signal currents of Q3
and Q4 are summed. For differential input signals, the signal currents of Q3 and Q4 are
equal and opposite. Thus, the sum is twice the individual signal currents. This
completes the differential to single ended conversion.
The open circuit signal voltage appearing at this point is given by the product of the
summed signal currents and the paralleled collector resistances of Q4 and Q6. Since the
collectors of Q4 and Q6 appear as high resistances to the signal current, the open circuit
voltage gain of this stage is very high.
It should be noted that the base current at the inputs is not zero and the effective
(differential) input impedance of a 741 is about 2 MΩ The offset null pins can be used
in conjunction with a potentiometer to remove any offset voltage that would exist at the
output of the op-amp when zero signal is applied between the inputs.
Class A gain stage
The section outlined in magenta is the class A gain stage. It consists of two NPN
transistors in a Darlington configuration and uses the output side of a current mirror as
its collector load to achieve high gain. The 30 pF capacitor provides frequency selective
negative feedback around the class A gain stage to stabilise the amplifier in feedback
configurations. This technique is called Miller compensation and functions in a similar
manner to an op-amp integrator circuit. It is also known as 'dominant pole
compensation' because it introduces a dominant pole (one which masks the effects of
other poles) into the open loop frequency response. This pole can be as low as 10 Hz in
a 741 amplifier and it introduces a −3 dB loss into the open loop response at this
frequency. This is done to achieve unconditional stability of the amplifier down to unity
closed loop gain using non-reactive feedback networks and makes this type of internally
compensated amplifier easier to use.
Output bias circuitry
The green outlined section (based around Q16) is a voltage level shifter or Vbe
multiplier; a type of voltage source. In the circuit as shown, Q16 provides a constant
voltage drop between its collector and emitter regardless of the current passing through
the circuit. If the base current to the transistor is assumed to be zero, and the voltage
between base and emitter (and across the 7.5 kΩ resistor) is 0.625 V (a typical value for
a BJT in the active region), then the current flowing through the 4.5 kΩ resistor will be
the same as that through the 7.5 kΩ, and will produce a voltage of 0.375 V across it.
This keeps the voltage across the transistor, and the two resistors at 0.625 + 0.375 = 1
V. This serves to bias the two output transistors slightly into conduction reducing
crossover distortion. In some discrete component amplifiers this function is achieved
with (usually 2) silicon diodes.
Output stage
The output stage (outlined in cyan) is a Class AB push-pull emitter follower (Q14, Q20)
amplifier with the bias set by the Vbe multiplier voltage source Q16 and its base
resistors. This stage is effectively driven by the collectors of Q13 and Q19. The output
range of the amplifier is about 1 volt less than the supply voltage, owing in part to
Vce(sat) of the output transistors.
The 25 Ω resistor in the output stage acts as a current sense to provide the output current
limiting function which limits the current flow in the emitter follower Q14 to about 25
mA for the 741. Current limiting for the negative output is done by sensing the voltage
across Q19's emitter resistor and using this to reduce the drive into Q15's base. Later
versions of this amplifier schematic may show a slightly different method of output
current limiting. The output resistance is not zero as it would be in an ideal op-amp but
with negative feedback it approaches zero.
Note: while the 741 was historically used in audio and other sensitive equipment, such
use is now rare because of the improved noise performance of more modern op-amps.
Apart from generating noticeable hiss, 741s and other older op-amps may have poor
common-mode rejection ratios and so will often introduce cable-borne mains hum and
other common-mode interference, such a switch 'clicks', into sensitive equipment.
Class A
Class A amplifiers amplify over the whole of the input cycle such that the output signal
is an exact scaled-up replica of the input with no clipping. Class A amplifiers are the
usual means of implementing small-signal amplifiers. They are not very efficient — a
theoretical maximum of 50% is obtainable, but for small signals, this waste of power
is still extremely small, and can be easily tolerated. Only when we need to create output
powers with appreciable levels of voltage and current does Class A become
problematic. In a Class A circuit, the amplifying element is biased such that the device
is always conducting to some extent, and is operated over the most linear portion of its
characteristic curve (known as its transfer characteristic or transconductance curve).
Because the device is always conducting, even if there is no input at all, power is
wasted. This is the reason for its inefficiency.
Class A Amplifier
If high output powers are needed from a Class A circuit, the power wastage will become
significant. For every watt delivered to the load, the amplifier itself will, at best, waste
another watt. For large powers this will call for a large power supply and large heat sink
to carry away the waste heat. Class A designs have largely been superseded for audio
power amplifiers, though some audiophiles believe that Class A gives the best sound
quality, due to it being operated in as linear a manner as possible. In addition, some
aficionados prefer vacuum tube (or "valve") designs over transistors, for a number of
reasons: Tubes are more commonly used in class A designs, which have an
asymmetrical transfer function. This means that distortion of a sine wave creates both
odd- and even-numbered harmonics. They claim that this sounds more "musical" than
the purely odd harmonics produced by a symmetrical push-pull amplifier.[1][2] Though
good amplifier design can avoid inducing any harmonic patterns in a sound
reproduction system, the differences in harmonic content are essential to the sound of
intentional electric guitar distortion. Another is that valves use many more electrons at
once than a transistor, and so statistical effects lead to a "smoother" approximation of
the true waveform — see shot noise for more on this. Field-effect transistors have
similar characteristics to valves, so these are found more often in high quality amplifiers
than bipolar transistors. Historically, valve amplifiers often used a Class A power
amplifier simply because valves are large and expensive; Many Class A design uses
only a single device. Transistors are much cheaper, and so more elaborate designs that
give greater efficiency but use more parts are still cost effective. A classic application
for a pair of class A devices is the long-tailed pair, which is exceptionally linear, and
forms the basis of many more complex circuits, including many audio amplifiers and
almost all op-amps.
Class B and AB
Class B amplifiers only amplify half of the input wave cycle. As such they create a large
amount of distortion, but their efficiency is greatly improved and is much better than
Class A. Class B has a maximum theoretical efficiency of 78.5%. This is because the
amplifying element is switched off altogether half of the time, and so cannot dissipate
power. A single Class B element is rarely found in practice, though it can be used in RF
power amplifiers where the distortion is unimportant. However Class C is more
commonly used for this.
Class B Amplifier
A practical circuit using Class B elements is the complementary pair or "push-pull"
arrangement. Here, complementary devices are used to each amplify the opposite halves
of the input signal, which is then recombined at the output. This arrangement gives
excellent efficiency, but can suffer from the drawback that there is a small glitch at the
"joins" between the two halves of the signal. This is called crossover distortion. A
solution to this is to bias the devices just on, rather than off altogether when they are not
in use. This is called Class AB operation. Each device is operated in a non-linear region
which is only linear over half the waveform, but still conducts a small amount on the
other half. Such a circuit behaves as a class A amplifier in the region where both
devices are in the linear region, however the circuit cannot strictly be called class A if
the signal passes outside this region, since beyond that point only one device will
remain in its linear region and the transients typical of class B operation will occur. The
result is that when the two halves are combined, the crossover is greatly minimised or
eliminated altogether.
However, it is important to note that while the efficiency of Class AB is greater than
Class A, it is less than Class B.
Class B Push-Pull Amplifier
Class B or AB push-pull circuits are the most common form of design found in audio
power amplifiers. Class AB is widely considered a good compromise for audio
amplifiers, since much of the time the music is quiet enough that the signal stays in the
"class A" region, where it is reproduced with good fidelity, and by definition if passing
out of this region, is large enough that the distortion products typical of class B are
relatively small. Class B and AB amplifiers are sometimes used for RF linear amplifiers
as well.
Negative feedback
Feedback feeds the difference of the input and part of the output back to the input in a
way that cancels out part of the input. The main effect is to reduce the overall gain of
the system. However the unwanted signals introduced by the amplifier are also fed
back. Since they are not part of the original input, they are added to the input in opposite
phase, subtracting them from the input.
Careful design of each stage of an open loop (non-feedback) amplifier can achieve
about 1% distortion. With negative feedback, 0.001% is typical. Noise, even crossover
distortion can be practically eliminated. Feedback was originally invented so that
replacing a burnt-out vacuum tube would not change an amplifier's performance
(manufacturing realities require that tubes and transistors with the same part number
will have close but not identical gain). Negative feedback also compensates for
changing temperatures, and degrading or non-linear components. While amplifying
devices can be treated as linear over some portion of their characteristic curve, they are
inherently non-linear; their physics dictates that they operate using a square law. The
result of non-linearity is distortion.
The application dictates how much distortion a design can tolerate. For hi-fi audio
applications, instrumentation amplifiers and the like, distortion must be minimal, often
better than 1%.
While feedback seems like a universal fix for all the problems of an amplifier, many
believe that negative feedback is a bad thing. Since it uses a loop, it takes a finite time
to react to an input signal, and for this short period the amplifier is "out of control." A
musical transient whose timing is of the same order as this period will be grossly
distorted, even though the amplifier will show incredibly good distortion performance
on steady-state signals. This, essentially, is the rationale for the existence of "transient
intermodulation distortion" in amplifiers which was exhaustively discussed and debated
from the late 1970s through much of the 1980s [3]. Proponents of feedback refute this,
saying that the feedback "delay" is of such a short order that it represents a frequency
vastly outside the bandwidth of the system, and such effects are not only inaudible, but
not even present, as the amplifier will not respond to such high frequency signals.
The argument has caused controversy for many years, and has led to all sorts of
interesting designs — such as feedforward amplifiers (e.g. digital signals on many cellsite base-station transmitters are precompensated for the radio amplifier's distortion).
The fact remains that the majority of modern amplifiers use considerable amounts of
feedback, though many of the high-end audiophile designs seek to minimise this.
Whatever the merits of these arguments about its effect on waveform distortion,
feedback also affects the output impedance of the amplifier and therefore its damping
factor. Roughly speaking, the damping factor is a measure of the ability of the amplifier
to control the speaker. All other things being equal, the greater the amount of feedback,
the lower its output impedance and the higher its damping factor. This has an effect on
the low frequency performance of many speaker systems where low damping factors
lead to irregular bass response.
The concept of feedback is used in operational amplifiers to precisely define gain,
bandwidth and other parameters.
A practical circuit
For the purposes of illustration, this practical amplifier circuit is described. It could be
the basis for a moderate-power audio amplifier. It features a typical (though
substantially simplified) design as found in modern amplifiers, with a class AB pushpull output stage, and uses some overall negative feedback. Bipolar transistors are
shown, but this design would also be realisable with FETs or valves.
A practical amplifier circuit
The input signal is coupled through capacitor C1 to the base of transistor Q1. The
capacitor allows the AC signal to pass, but blocks the DC bias voltage established by
resistors R1 and R2 so that any preceding circuit is not affected by it. Q1 and Q2 form a
differential amplifier (an amplifier that multiplies the difference between two inputs by
some constant), in an arrangement known as a long-tailed pair. This arrangement is used
to conveniently allow the use of negative feedback, which is fed from the output to Q2
via R7 and R8. The negative feedback into the difference amplifier allows the amplifier
to compare the input to the actual output. The amplified signal from Q1 is directly fed to
the second stage, Q3, which provides further amplification of the signal, and the DC
bias for the output stages, Q4 and Q5. R6 provides the load for Q3 (A better design
would probably use some form of active load here, such as a constant-current sink). So
far, all of the amplifier is operating in Class A. The output pair are arranged in Class AB
push-pull, also called a complementary pair. They provide the majority of the current
amplification and directly drive the load, connected via d.c. blocking capacitor C2. The
diodes D1 and D2 provide a small amount of constant voltage bias for the output pair,
just biasing them into the conducting state so that crossover distortion is minimised.
This design is simple, but a good basis for a practical design because it automatically
stabilises its operating point, since feedback internally operates from DC up through the
audio range and beyond. Further circuit elements would probably be found in a real
design that would roll off the frequency response above the needed range to prevent the
possibility of unwanted oscillation. Also, the use of fixed diode bias as shown here can
cause problems if the diodes are not both electrically and thermally matched to the
output transistors — if the output transistors turn on too much, they can easily overheat
and destroy themselves, as the full current from the power supply is not limited at this
stage. A common solution to help stabilise the output devices is to include some emitter
resistors, typically an ohm or so. Calculating the values of the circuit's resistors and
capacitors is done based on the components employed and the intended use of the amp.
For the basics of radio frequency amplifers using valves, see Valved RF amplifiers.
Class C
Class C amplifiers conduct less than 50% of the input signal and the distortion at the
output is high, but efficiencies of up to 90% can be reached. Some applications can
tolerate the distortion, such as audio bullhorns. A much more common application for
Class C amplifiers is in RF transmitters, where the distortion can be vastly reduced by
using tuned loads on the amplifier stage. The input signal is used to roughly switch the
amplifying device on and off, which causes pulses of current to flow through a tuned
circuit. The tuned circuit will only resonate at particular frequencies, and so the
unwanted frequencies are dramatically suppressed, and the wanted full signal (sine
wave) will be abstracted by the tuned load. Provided the transmitter is not required to
operate over a very wide band of frequencies, this arrangement works extremely well.
Other residual harmonics can be removed using a filter.
Class C Amplifier
Crossover distortion
The term crossover signifies the "crossing over" of the signal from the upper transistor
to the lower and vice-versa. Crossover distortion can be suppressed by using a slight
forward bias in the base circuit such that the transistors are idling at a small output
current and also by the use of negative feedback. The forward bias causes the circuit to
operate in class-AB mode.
Most modern power amplifiers (including those used in hi-fi) employ class-AB in their
output stages as it offers good to very good efficiency and distortion figures.
The next image shows a typical class-B emitter-follower complementary output stage.
Most of the accompanying circuitry has been omitted for clarity. Under no signal
conditions, the output is exactly mid-way between the supplies (i.e., at 0V). When this
is the case, the base-emitter bias (voltage) available both transistors is zero and so they
are in the cut-off region, i.e., the transistors are not conducting.
Consider a positive going swing: As long as the input is less than the required forward
VBE drop (≈ 0.65V) of the upper NPN transistor, it will remain off or conduct very little
- this is the same as a diode operation as far as the base circuit is concerned, and output
voltage does not follow the input (the lower PNP transistor is still off because its baseemitter diode is being reverse biased by the positive going input). The same applies for
the lower transistor, but for a negative going input. Thus, between about ±0.65V of
input, the output voltage is not a true replica or amplified version of the input and we
can see that as a "kink" in the output waveform near 0V (or where one transistor stops
conducting and the other starts). This kink is known as crossover distortion and it
becomes more evident and intrusive when the output voltage swing is reduced.
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